Fuel Cell 10 kW Inverter System
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2003 International Future Energy Challenge Competition Final Report A 10KW Fuel Cell Inverter System Submitted by Seoul National University of Technology Student Team Faculty Advisor Dr. Sewan Choi May 18, 2003
description
Low cost inverter system
Transcript of Fuel Cell 10 kW Inverter System
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2003 International Future Energy Challenge Competition
Final Report
A 10KW Fuel Cell Inverter System
Submitted by
Seoul National University of Technology
Student Team
Faculty Advisor
Dr. Sewan Choi
May 18, 2003
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TABLE OF CONTENTS
1. Introduction 2 2. Management 3
2.1 Team Organization 3 2.2 Education Impact 4
2.3 Project time line 5 2.4 Project Budget 5 3. Topology Evaluation 6 3.1 Two Topologies 6 3.2 Power Component Design 8 3.3 Cost Evaluation 9 3.4 Efficiency Evaluation 12 4. Design Rationale 14 4.1 Front end DC-DC Converter 14 4.2 Inverter 21 4.3 Bi-directional DC-DC Converter 28
4.4 System Interface 35 4.5 Heat Sink and Packaging 37
5. Simulation 41 6. Experimental Result 43 7. Performance Evaluation 46 8. Bill of materials 47 9. Cost Analysis 47 10. Conclusion 49 11. Reference 50
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List of Figures
Fig. 2.1 SNUT student team Organization Chart 4 Fig. 3.1 Proposed power circuit topologies 7 Fig. 4.1 Block diagram of the SNUT fuel cell inverters systems 14 Fig. 4.2 Circuit diagram of front end DC-DC converters 15 Fig. 4.3 Main waveforms of the front end DC-DC converters 15 Fig. 4.4 Control block diagram for front end DC-DC converters 20 Fig. 4.5 Circuit diagram of the inverter 21 Fig. 4.6 Equivalent circuit for a LC output filter 24 Fig. 4.7 Equivalent circuit for a non-linear load 25 Fig. 4.8 Control method of balancing the capacitor voltages 27 Fig. 4.9 Bi-directional DC-DC converter 29 Fig. 4.10 Control block diagram for the bi-directional
DC-DC converter 30
Fig. 4.11 Inductor voltage & current waveforms 31 Fig. 4.12 Display of RS-232 36 Fig. 4.13 Thermal equivalent circuit 39 Fig. 4.14 Heat sink for front-end DC-DC converter 40 Fig. 4.15 Heat sink for inverter and bi-directional converter 41 Fig. 5.1 Simulated waveforms 42 Fig. 6.1 Experimental waveforms (4.4kW load) 44 Fig. 6.2 Experimental waveforms ( 2KW 2.7KW ) 45 Fig. 6.3 Photograph of the SNUT fuel cell Inverter 46
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List of Tables
Table 2.1 Project budget 6 Table 3.1 System parameters for power circuit design 9 Table 3.2 Component ratings of Scheme I 10 Table 3.3 Component ratings of Scheme II 11 Table 3.4 Cost estimates 12 Table 3.5 Power loss estimates 13 Table 4.1 47054-EC Magnetic data 19 Table 4.2 Voltage and current ratings of the switch 34 Table 4.3 Power dissipation in the device used 40 Table 4.4 Thermal characteristics for the heat sink design 40 Table 7.1 Experimental performance (no load to 4.4kW load) 47 Table 8.1 Bill of Materials 48 Table 9.1 Cost spread sheet for front end DC-DC converter 48 Table 9.2 Cost spread sheet for inverter 49 Table 9.3 Cost spread sheet for bi-directional converter 49
Appendices
Appendix 51
A.1 : Schematic for the sensing board
A.2 : Schematic for the sensing and protection
A.3 : DSP board
A.4 : Inverter gate driver
B : Project time line
C : Transformer core selection by area product distribution
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SNUT Fuel Cell Inverter Team
Student Members
Minsoo Jang Min Koo
Jaehyuck Jung Sangmin Jung
Taehoon Kim Jinwook Oh
Hyunjung Kim Namki Lee
Jinhee Lee Joonseo Lee
Jinsang Jo Kangsuk Lee
Minkook Kim Byungsoo Nho
Seungjoo Cheon
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Summary
The objective of the 2003 Future Energy Challenge competition is to develop a low cost
10kW power processing unit for a fuel cell system. The SNUT team, which is composed of
senior undergraduate students, graduate students and faculty advisors, has been launched for the
competition.
This report discusses the power circuit topologies for the SNUT inverter system by evaluating
the topologies in a practical way. After researching several topologies a topology is chosen and
the component ratings are designed along with through analysis on the chosen topology. The
simulation is performed to verify the design and control of the proposed topology. A hardware
prototype capable of supplying 10kW load was built and tested in the laboratory of SNUT.
Experimental performances on some design items are compared to minimum target
requirements of the inverter system. The cost analysis is done based on the spreadsheets
evaluation forms provided in the 2003 FEC workshop. Some conclusions are made to meet the
minimum target requirement in the final competition.
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1. Introduction
The environmental concern is now the driving force for alternative energy. Fuel cell power
generation systems are expected to see increasing practical use due to the several advantages
over conventional generation systems. These advantages include 1) low environmental pollution
2) highly efficient power generation 3) diversity of fuels(natural gas, LPG, methanol and
naphta) 4 ) reusability of exhaust heat 5) modularity and 6) faster installation [1]. Fuel cells are
generally characterized by the type of electrolyte that they use. Solid oxide fuel cells (SOFC)
have grown in recognition as a viable high temperature fuel cell technology. The most striking
quality of SOFCs is that the electrolyte is in solid state and is not a liquid electrolyte. The high
operating temperature up to 1000C allows internal reforming, promotes rapid kinetics with
non-precious materials and produces high quality byproduct heat for cogeneration or for use in a
bottoming cycle. A number of different fuels can be used from pure hydrogen to methane and
carbon monoxide. The major advantage of SOFC lies in its efficiencies raging from 55 to 60%
[2].
In general, the function of a power conditioning system in a fuel cell generation system is to
convert the DC output power from the fuel cell to regulated AC power. There may be two stages
of power converters. A DC-DC converter converts the low voltage DC output from the fuel cell
to a level at which an inverter can safely operate. The inverter is used to invert the DC output
from the DC-DC converter to a suitable AC voltage. The power conditioning unit that basically
consists of an inverter is required to have the following characteristics: 1) allowable for wide
output voltage regulation of fuel cell 2) controllability of output voltage 3) available for isolated
operation and line parallel operation 4) fast reactive power dispatch 5) low output harmonics 6)
high efficiency and 7) suitable for high power system [3]. Fuel cell production costs are
currently decreasing and have nearly achieved energy costs that are competitive with local
utility rates. The inverter cost must also decrease while at the same time increasing efficiency,
reliability, and power quality levels. The cost reduction of the power processing unit will enable
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the fuel cell systems to penetrate rapidly into the utility market.
The objective of the Fuel Cell Inverter Challenge is to develop a 10kW low-cost power
processing systems that support the commercialization of a SOFC power generation system to
provide non-utility and ultra-clean residential electricity. The target cost of the stand-alone
10kW power processing unit will be less than $40/kW in high volume. Further, emphasis will
also be placed on high energy efficiency as this has direct impact of size and cost of the SOFC
system and overall system fuel efficiency. Another key objective of this competition is to
promote design education in the undergraduate curriculum at Seoul National University of
Technology in conjunction with faculties and industry experts in the power electronics and
energy conversion area.
2. Management
2.1 Team Organization
The Seoul National University of Technology (SNUT) has formed a multi-disciplinary team
consisting of nine undergraduate and six graduate(master) students and three faculty advisors.
The undergraduate students from Department of Control and Instrumentation Engineering(CIE),
Department of Electrical Engineering(EE), and Department of Mechanical Design(MD) have
been selected by public notice in our university. The students are divided into six sub-teams (1)
Inverter and DSP (2) Front-end DC-DC converter (3) Bi-directional DC-DC converter and
Battery control (4) Sensors and Protection (5) System integration and Interface (6) Heat sink
and Packaging. The current organizational chart is shown in Fig. 2.1. This interdisciplinary
approach will allow the team to address thermal management, packaging and case issues.
Dr. Sewan Choi, professor in the power electronics area, served as the lead Faculty Advisor
for the team. Other faculty advisors to this project include: Dr. Kiyong Kim with expertise in
control systems and Dr. Youngseog Kim, professor in the mechanical design area, with expertise
in heat sink design and packaging.
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Inverter & DSP Minsoo Jang (CIE, G) Jinwook Oh (CIE, UG)
Front-end DC-DC converter Jinhee Lee (CIE, G)
Namki Lee (CIE, UG)
Bi-directional DC-DC converter and Battery control Jaehyuck Jung (CIE, G)
Jinsang Jo (CIE, G) Kangsuk Lee (EE, UG)
Faculty Advisors
Dr. Sewan Choi (CIE)
Dr. Kiyong Kim (CIE)
Dr. Youngseog Kim (MD)
Sensors and Protection Hyunjung Kim (CIE, G) Taehoon Kim (CIE, G)
System integration and Interface Min Koo (CIE, UG)
Sangmin Jung (CIE, UG) Joonseo Lee (CIE, UG)
Heat sink and Packaging Minkook Kim (CIE, UG) Byungsoo Nho (MD, UG)
Seungjoo Cheon (CIE, UG)
Fig. 2.1 SNUT team Organization Chart
2.2 Educational Impact
The undergraduate students in most departments in our university are required to take Design
Project Courses. For example, the students in the Department of Control and Instrumentation
Engineering must take Design Project I(CIE322) and Design Project II(CIE415) through the fall
semester of junior year and the spring semester of senior year. Throughout the courses, the
undergraduate students choose their own topic, do some research on the topic, design and build
the hardware prototype. In the department of CIE, as a mandatory graduation requirement, all
the senior students should submit a written report, present what they have worked on with
working prototype and be judged by the committee from faculty members. The students have
to take all the courses necessary to complete the project. The 2003 Future Energy Challenge
provided a good topic for the undergraduate students to participate in the competition. All the
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undergraduate students on the team will receive credits in Design Project courses(CIE322 and
CIE415).
The SNUT team has been holding weekly meetings with all of its members to discuss the
project. We have had a series of technical seminars mostly by ourselves to get the practical
background on design of the fuel cell power processing system. Each of graduate student led the
seminar on some specific topic. Some experts from industries had been sometimes invited to
give students a practical knowledge.
These kinds of efforts have been shown to be successful in attracting students. In fact, in the
spring semester of 2003 two student members, Jinhee Lee and Jinsang Jo, joined the graduate
program in power electronics of SNUT and continue participating this 2003 FEC competition.
In the spring semester of 2004 four student members will apply for the graduate program in
power electronics.
2.3 Project Timeline
The project time line for the SNUT team is sown in Appendix B. The team plans to
construct the prototype in two steps ; the preliminary prototype by the mid of November 2002
and the final prototype by the end of February 2003. In November 2002, the team submitted the
written report and presented working prototype as a graduation requirement. The preliminary
prototype was evaluated by peer reviews from faculty advisors and industry experts. The team
incorporated feedback from reviewers and any design changes to final prototype.
2.4 Project Budget
The project budget for the SNUT Future Energy Challenge team is shown in Table 2.1. The
budget only takes into account parts for the power processing system and travel for fundraising
and the competition excluding any labor and equipment. The undergraduate students take part in
the competition as a choice of topic for the Design Project course and the graduation
requirement. Also, the department of CIE, SNUT already has laboratory facilities equipped with
many power electronics instruments and equipments.
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Table 2.1 Project budget for the SNUT Future Energy Challenge team
Classification Amount
Power Device (IGBT, MOSFET, DIODE) $ 2,000
Battery $ 700
DSP Evaluation $ 500
PCB $ 1,500
Sensors $ 700
Inductors $ 800
Capacitors $ 500
Transformer $ 2,000
Analog & Digital Ics $ 1,000
Parts
Cable, bus bar and case $ 2,000
Company Presentations $ 1,000
Work shop $ 5,000 Travel
Final Competition $ 10,000
Copies $ 300 Miscellaneous
Lab supplies $ 500
Sub Total $ 28,000
Support from our university and industries is essential to the teams successful project
performance not only for financial sponsorship, but also for industrial experience. The SNUT
team faculty advisors and students have been trying to secure the necessary funding for this
project from the school, industries including national laboratories. The SNUT team secured the
sponsorship and commitment from the school and some industries, and has been trying solicit
additional support for the proposed project.
3. Topology Evaluation
3.1 Two Topologies
The SNUT team decided to use the low voltage (48V) battery, which is supposed to be
provided at the competition test site, as a secondary energy storage to supply transient loads.
The SNUT team considered two types of power circuit topologies for the SOFC power
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processing system : Scheme I shown in Fig. 3.1(a) and Scheme II shown in Fig. 3.1(b). In
Scheme I, as shown in Fig. 3.1(a), the DC voltage from the fuel cell, 29VDC nominal, is first
boosted to 48V via a non-isolated boost converter. The 48V battery bank in the fuel cell system
is connected to the 48V DC link so that power flow to and from the battery is controlled by the
current control of the boost converter. The 48VDC from the boost converter is then converted to
400VDC via an isolated high frequency DC-DC converter. The high frequency DC-DC converter
could be push-pull, half-bridge or full-bridge types. The full-bridge type is considered suitable
for 10KW of high frequency DC-DC conversion.
Battery48V
C1
T1
D4
D2S6 S7
C2
C3
S9S8
L2
C4
L3
D5
Fuel Cell22 ~ 41V
S1
L1D1
L4
C5
L5
A
N
D3S2 S3
S4 S5
B
120Vac60HZ
120Vac60HZ
240Vac60HZ
+
-
Vdc400v
(a) Scheme I
L5
L4
120Vac60HZ
120Vac60HZ
120Vac60HZ
A
B
N
+
-
Vdc400V
T2
S10S9Battery48V
L3
S7 S8
S6S5
T1
C inL2
L1
C2
C1
D5 D6
D8D7
D4D3
D1 D2
S4S3
S2S1
Fuel Cell22 ~ 41 V
C3
C4
S14S13
S12S11
(b) Scheme II
Fig. 3.1 Proposed power circuit topologies
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The DC-DC conversion stage includes a transformer isolation for safety, protection and to
meet the stringent FCC Class-A standards. The 400V DC-DC converter output is then converted
to 120V/240V, 50/60 Hz, single-phase AC by means of a PWM inverter stage. An output L-C
filter stage is employed to reduce the ripple component and draw a low THD AC waveform.
Fig.3.1(b) shows an alternative power circuit configuration (Scheme II) for the SOFC inverter
system. The DC voltage from the fuel cell , 29VDC nominal, is first converted to 400VDC via an
isolated high frequency DC-DC converter. The high frequency DC-DC converter could be push-
pull, half-bridge or full-bridge types. The full-bridge type with two diode bridges connected
in series at the secondary was chosen as a 5KW of front-end DC-DC conversion. The DC-
DC conversion stage also includes a transformer isolation for safety, protection and to meet the
stringent FCC Class-A standards as well. The 48V battery bank in the fuel cell system is
connected to the 400V DC link via a bi-directional DC-DC converter which is also operated at
high frequency. The low voltage(48VDC) battery side and the high voltage(400VDC) dc link side
of the bi-directional DC-DC converter could be current-source full-bridge type and voltage-
source full-bridge type, respectively, or vice versa. A current-sourced push-pull type using
MOSFETs and a voltage-sourced full-bridge type using IGBTs were chosen at the battery
side and the dc link side, respectively. The 400V DC-DC converter output is then converted to
120V/240V, 50/60 Hz, single-phase AC by means of a PWM inverter stage. An output L-C filter
stage is employed to reduce the ripple component and draw a low THD AC waveform.
The common topology chosen for both of the schemes to provide the split phase output
was two half-bridge inverters. Both of the topologies have thoroughly been examined from
cost and efficiency standpoint, and one of them was adopted for this project.
3.2 Power Component Design
In this section power components of the two schemes are designed so that the designed
values are used to compare cost and efficiency of the two schemes. The output displacement
factor is assumed to be unity for design of the fuel cell power processing system rated at
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10KW. Table 3.1 shows some system parameters for power circuit design of the two
schemes.
Table 3.2 and 3.3 list the designed ratings of the power components in the two schemes
using system parameters in Table 3.1. Based on the designed values, actual devices were
selected from some manufactures. Appropriate safety margins were considered for actual
device selection. The design procedure and device selection presented here is not unique,
but they could be used to relatively compare both of the schemes from cost and efficiency
standpoint.
3.3 Cost Evaluation
In this section the two schemes are compared each other from a cost standpoint. The
spreadsheets evaluation forms presented at the 2003 FEC workshop is used to perform
relative cost estimates. Cost factors have been obtained based on the designed vales, shown
in Table 3.2 and 3.3, without safety margin. The resulting cost estimates of the two schemes
is shown in Table 3.4. It can be noticed from Table 3.4 that power switches, capacitors,
transformers employed in both schemes did not give much difference in cost. However, the cost
of the inductor in the non-isolate boost converter section of Scheme I is significant due to its
high current capacity. Therefore, it can be concluded that Scheme II is superior to Scheme I in
cost.
Table 3.1 System parameters for power circuit design of the two schemes.
Scheme I Scheme II
Non-isolated Boost
Isolated DC-DC
Inverter Front-endDC-DC
Inverter Bi-directional
DC-DC Switching frequency
40kHz 20kHz 20kHz 25kHz 20kHz 20kHz
Input voltage
22 ~ 41V 42 ~ 57.6V 400V 22 ~ 41V 400V 42 ~ 57.6V
Output voltage
42 ~ 57.6V 400V 60Hz 120VAC(Split phase)
400V 60Hz 120VAC (Split phase)
400V
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Table 3.2 Component ratings of Scheme I
Section Component Designed value Actual device Selection
Vpeak (V) 57.6 MOSFET (S1) Irms (A) 214.4
APT APT10M07JVR
(100V, 225A, 7m)
Vpeak (V) 57.6 Diode (D1) Irms (A) 266.2
IXYS DSEI2x161-12P
(1200V, 2X128A, trr = 35ns)
Inductance 50 uH Inductor (L1) Irms (A) 272.5
MAGNETICS 43208 (EI)
Capacitance 3300uF
Non -isolated
Boost
Capacitor (C1) Vpeak (V) 77.7
Samwha SZ2A338M35100
(100V, 3300uF, ESR 0.08)
Vpeak (V) 57.6 MOSFET (S2, S3, S4, S5) Irms (A) 336.66
IXYS IXFN340N07
(70V, 340A, 4m)
Vrms (V) 47.3 Transformer (T1) Irms (A) 389.8
MAGNETICS 49925 (U)
Vpeak (V) 410 Diode (D2 ~ D9) Irms (A) 13.75
IXYS DESI 30-10A
(1000V, 30A, trr = 50ns)
Inductance 100 uH Inductor (L2, L3) Ipeak (A) 27.3
Chang-sung CH270125E (Toroid)
Capacitance 5000 uF
Isolated DC-DC
Capacitor (C2, C3) Vpeak (V) 210
Samwha GF2G688M76160
(400V, 6800uF, ESR 0.04)
Vpeak (V) 420 IGBT (S6, S7, S8, S9) Irms (A) 50
TOSHIBA MG50Q2YS50
(600V, 50A, VCE(sat) 2.7V)
Inductance 93 uH Inductor (L4, L5) Irms (A) 42A
Chang-sung CH572060E (Toroid)
Capacitance 16uF
Inverter
Capacitor (C4, C5) Vpeak (V) 170V
Digital Tech
(250V, 15uF, ESR 0.04)
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Table 3.3 Component ratings of Scheme II
Section Component Designed value Actual Device Selection
Vpeak (V) 41 MOSFET (S1, S2, S3, S4)
Irms (A) 177.4
IXYS IXFN340N07
(70V, 340A, 4m)
Vrms (V) 47.3 Transformer (T1)
Irms (A) 389.8
MAGNETICS 49925 (U)
Vpeak (V) 410 Diode (D1 D8)
Irms (A) 8.8
IXYS DESI 30-10A
(1000V, 30A, trr = 50ns)
Inductance 100 uHInductor (L1, L2)
Irms (A) 12.6
Chang-sung CH270125E (Toroid)
Capacitance 5000 uF
Front-end DC-DC
Capacitor (C1, C2)
Vpeak (V) 210
Samwha GF2G688M76160
(400V,6800uF, ESR 0.04)
Vpeak (V) 420 IGBT (S5, S6, S7, S8)
Irms (A) 50
TOSHIBA MG50Q2YS50
(600V, 50A, VCE(sat) 2.7V)
Inductance 93 uH Inductor (L3, L4)
Irms (A) 42A
Chang-sung CH572060E (Toroid)
Capacitance 16uF
Inverter
Capacitor (C3, C4)
Vpeak (V) 170V
Digital Tech
(250V, 15uF, ESR 0.04)
Vpeak (V) 140 MOSFET (S9, S10) Irms (A) 77.5
APT APT20M20JFLL
(200V, 104A, 20m)
Vrms (V) 58 Transformer (T2)
Irms (A) 73.3
MAGNETICS 49925 (U)
Inductance 40 uH Inductor (L5)
Irms (A) 113
Chang-sung CH572060E (Toroid)
Vpeak (V) 420
Bi - Directional
DC-DC
IGBT (S11,S12, S13,S14)
Irms (A) 13.1
TOSHIBA MG50J2YS50
(600V, 50A, VCE(sat) 2.7V)
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Table 3.4 Cost estimates of the two schemes according to FEC 2003 cost spreadsheet
Scheme I Scheme II Component
Desig. Qty Cost
Factor Unit
Cost ($)ExtendedCost ($)
Desig. QtyCost
FactorUnit
Cost ($) ExtendedCost ($)
S1 1 12349 10,44 10.44 S1 ~ S4 4 7273.4 7.71 30.85 MOSFET S2 ~ S5 4 22419 14.75 50.01 S9, S10 2 10850 9.64 19.27
D1 1 15333 3.45 3.45 D1 ~ D8 8 3608 2.38 9.55 Diode
D2 ~ D5 4 5637 2.57 10.30
S5 ~ S9 4 21000 8.43 33.7 S5 ~ S8 4 21000 8.45 33.7 IGBT
S11~S14 4 5502 2.3 9.2
T1 1 18437 26.55 26.55 T1 1 18437 24.7 24.7 Transformer
T2 1 4251.4 10.94 10.94
L1 1 3.69 249.51 249.51 L1, L2 2 0.01 41.8 83.7
L2, L3 2 0.07 44.5 89 L4, L5 2 0.16 49.81 99.62 Inductor
L4, L5 2 0.16 49.81 99.62 L5 1 0.5 68.7 68.7
C1 1 19.92 2.85 2.85 C1, C2 2 220 30.63 61.27
C2, C3 2 220 30.63 61.26 C4, C5 2 0.46 0.16 0.32 Capacitor
C4, C5 2 0.46 0.16 0.32
Total Cost 636.97 451.82
3.4 Efficiency Evaluation
In this section, efficiency is compared each other by means of power loss calculation.
For reasonable efficiency comparison the power loss is calculated based on when the output
power is 5KW. That is, efficiency is evaluated at the steady-state load condition excluding
battery operation. As shown in Table 3.2 and 3.3, actual devices from some manufactures
have been selected with appropriate safety margin based on the designed value. From the
datasheet of the selected devices, the power loss can be calculated in the following manner.
The switching loss and conduction loss are considered for power switching devices such as
diodes, MOSFETs and IGBTs. The core loss and copper loss of the transformer can be
calculated from the selected core and wire from the manufacture. Only copper loss has been
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considered for inductor since copper loss dominates when an inductor is operated as a filter
at high frequency. Capacitor loss is calculated using ESR of the selected capacitor.
The efficiency of the both schemes can be obtained based on the power loss estimates in
Table 3.5. The estimated efficiencies are 85.4% for Scheme I and 94.6% for Scheme II,
respectively. Losses in the diode and the MOSFET in the non-isolate boost converter section of
Scheme I are significant. The losses in the two switches exceed 50% of the total loss of Scheme
I. Scheme II is superior to Scheme I in both cost and efficiency. In conclusion, the SNUT team
decided to choose Scheme II for the power circuit topology.
Table 3.5 Power loss estimates of the two schemes based on manufactures datasheet
Scheme I Scheme II Device
Desig. Loss ExtendedLoss (W)
Desig. Loss Extended Loss (W)
Conduction loss 335.6 Conduction loss 13.2 D1
Switching loss 0.875 D1 ~ D8
Switching loss 0.012
Conduction loss 20.62 Diode
D2 ~ D5 Switching loss 0.024
Conduction loss 122.1 Conduction loss 63.3 S1
Switching loss 40.69 S1 ~ S4
Switching loss 29.8
Conduction loss 63.3 MOSFET
S2 ~ S5 Switching loss 29.8
Conduction loss 36.7 Conduction loss 36.7 IGBT S6 ~S9
Switching loss 29.4 S5 ~ S8
Switching loss 29.4
Core loss 31.84 Core loss 15.9 Transformer T1
Copper loss 45.77 T1
Copper loss 24.87
L1 Copper loss 7.77 L1, L2 Copper loss 0.074
Core loss 1.65 L2, L3 Copper loss 0.148 L3, L4
Copper loss 0.007
Core loss 1.65
Inductor
L4, L5 Copper loss 0.007
C1 Capacitor loss 29.15 C1, C2 Capacitor loss 0.0036
C2, C3 Capacitor loss 0.15 C3, C4 Capacitor loss 69.2 Capacitor
C4, C5 Capacitor loss 69.2
Total Loss 864.8 284.12
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4. Design rationale
According to the topology evaluation in Section 3, the SNUT team decided to choose
Scheme II, shown in Fig. 3.1(b) , for the power circuit topology. The block diagram for the
SNUT fuel cell inverter system is shown in Fig. 4.1. The inverter system consists of a front
end DC-DC converter, a DC-AC inverter and a bi-directional DC-DC converter. Both the
fuel cell current control and the dc link voltage control are performed for the front end DC-
DC converter to improve dynamic performance of the system during a transient state.
The bi-directional DC-DC converter is operated to charge or discharge the battery according
to the current reference and the mode of operation determined by the DSP. The two PWM
controllers are employed for charge and discharge modes of operation, respectively.
Fuel Cell
22~41V
Front-endDC-DC Inverter
Battery
48V
Bi-directionalDC-DC
DSP320LF2407UCC 3895
UCC 3895&
UCC 3825
IBatt
VBatt
Vo
IoIdc
IFCVFC
Vdc
IBatt
+
-VBatt
VFC-
+IFC Idc
Vdc-
+Io
+- VoLoad
120V/240V60HZ
AC output
400V
Fig. 4.1 Block diagram of the SNUT fuel cell inverter system
4.1 Front end DC-DC converter
A Front end DC-DC converter is required to boost a unregulated fuel cell voltage of 29V
nominal to a regulated 400V, as shown in Fig. 4.2. The full-bridge type is a topology of
choice with which a phase-shift PWM technique can be implemented.
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Fuel Cell22 ~ 41 V
S1 S2
S3 S4
D2D1
D3 D4
D7 D8
D6D5
C1
C2
L1
L2C in
T1
+
+
-
-
VL
+
-
Vin
Vd
Vdc400V
+
-
+
- -
-+
+
Vsec1
Vsec2
Vpri1
IL Idc
Vpri2
Np : Ns
Fig. 4.2 Circuit diagram of front end DC-DC converter
The phase shift control can achieve zero voltage switching, reducing the losses in the
switch and therefore increasing system efficiency. High frequency transformers are
employed to allow a low voltage to be boosted to two split 200VDC buses for the DC link to
the Inverter. The two high frequency transformers connected in parallel supply two separate
diode bridges connected in series. The reason why two 2.5kW high frequency transformers
are employed instead of using a 5kW high frequency transformer is to reduce the leakage
inductances and therefore to reduce the duty loss.
Fig. 4.3 Main waveforms of the front-end DC-DC converter
-
16
The reduce duty loss also reduces turns ratio of the transformer. This reduces the voltage
rating of diodes in the secondary side and current rating of MOSFETs in the primary side.
Fig. 4.3 shows the main waveforms of the front-end DC-DC converter.
From the inductor voltage VL an equation can be written as,
=
2)1(
21
2)
21( DTVDTV
NNV dcdc
p
sin (1)
Therefore, the duty cycle of the proposed front-end DC-DC converter is obtained by,
ins
dcp
VNVN
D
=
4 (2)
According to eqn.(2), the duty cycle ranges 0.24 to 0.45 to regulate the dc link voltage of
400V when the fuel cell voltage varies between 22V and 41V.
4.1.1 Power component design
The power components of the front end DC-DC converter are designed in this section
with the following system parameters.
DC link power Pdc : 5kW
Switching frequency fs : 25kHz
Input voltage Vin: 22 ~ 41V
DC link voltage Vdc : 400V
Transformer turns ratio Np : Ns = 1 : 10
The maximum current at the dc link is,
AVPI
dc
dcdc 5.12== (3)
Filter inductor Design
During2
0 DTt
-
17
This inductance can be obtained by combining eqns.(2) and (4),
s
inp
s
fI
DVNND
L
=
)()21( (5)
Assuming a permissible ripple current on the inductor to be 50% of the maximum current
at the dc link or 6.25A peak to peak, the inductance can be calculated using eqn.(5) to be L
= 100 uH. The peak inductor current can also be calculated as,
AIII dcpeakL 6.1521
, =+= (6)
From Fig. 4.3, the rms value of inductor current can also be calculated as,
AI rmsL 6.12, =
Power switch design
The voltage rating of diodes and MOSFETs are calculated based on the moment at which
the fuel cell outputs a maximum of 41V at a minimum duty cycle of D = 0.24. The current
rating of diodes and MOSFETs are calculated based on the moment at which the fuel cell
draws a maximum current of 275A at a maximum duty cycle of D = 0.45.
a. Diodes
When the fuel cell voltage is 41V, the secondary winding voltage becomes 410V,
therefore, the peak voltage of a rectifier diode is 410V. A safety margin should be
considered due to the ringing phenomenon at the secondary winding of the high frequency
transformer. The voltage rating of the diode is determined to be 600V. A ultra-fast recovery
diode is chosen to lower the switching loss due to the high switching frequency
operation(25kHz). As the diode always conducts half of a switching cycle, the average
current rating of the diode can be obtained by,
AII dcavD 25.621
, == (7)
The peak current of the diode is identical to the peak current of the inductor. Then,
AII peakLpeakD 6.15,, == (8)
-
18
A ultra-fast recovery diode with a rating of 600V, 20A was selected from a manufacture.
b. MOSFETs
Power MOSFETs were selected as a switching device for the front-end DC-DC converter
since they should operate under the low voltage and high current condition. The peak
voltage of a MOSFET is 41V which is the maximum fuel cell voltage. Considering a safety
margin due to voltage spikes originated from the leakage inductance of the high frequency
transformer, the device voltage rating over 70V should be acceptable. By multiplying the av
and peak value of the diode by the turns ratio of the transformer, the av and peak values of
the MOSFET can be calculated be 125A and 275A, respectively. Considering safety
margins, MOSFETs with ratings of 70V, 340A were selected from a manufacture.
Transformer Design
As mentioned at the above, two 2.5KW high frequency transformers are employed and the
transformer design for the 2.5KW transformer is described in the following.
a. Core material
Ferrite is an ideal core material for transformers and inductors in the frequency range 20 KHz
to 3MHz, due to the combination of low core cost and core losses. Ferrite core is chosen as a
material of a high frequency transformer.
b. Core size
The power handling capacity of a transformer core can be determined by its area product
WaAc, where Wa is the available core window area, and Ac is the effective core cross-sectional
area. The SNUT team would follow the procedure for transformer core size selection provided
by Magnetics Co [8].
The area product is given by,
KfBeCPAW
s
dcca
=
4108 (9)
where Pdc is output power, C is current capacity, e is transformer efficiency, B is flux density,
-
19
Table 4.1 47054-EC Magnetic data (MAGNETICS Inc.)
AL values (mH/1000turns)
Materials Combination 2300
K (min.)
2500 R
(min.)
3000P
25%
5000F
(min.)
Ie
(cm)
Ae
(cm2)
MinimumArea
(cm2)
Ve
(cm3)
SET NOM.
Wt.
(gms)
WaAc (cm4)
EER - 2,440 2,650 4,240 23.1 3.39 3.14 78.6 396 34
fs is switching frequency and K is winding factor. The core type of choice is EER core, then
AmpcmC /1007.5 23= (10)
The transformer efficiency is assumed to be 90%. The winding factor is K = 0.3 (primary side
only). The flux density B is assumed be 4500(gauss). Then, the WaAc product is calculated as,
447.23 cmAW ca = (11)
Using the core selection table by area product distribution (refer to Appendix C), the core of
47054-EC was selected. Magnetic data for the selected core is shown in Table 4.1.
c. Number of turns
Once a core is chosen, the calculation of primary and secondary turns and their wire sizes are
readily accomplished. The number of primary turns is given by [8],
turnsfAB
VNs
Pp 02.62500039.320004
10414
10 88=
=
= (12)
Here, Vp is the peak primary voltage is and A is the cross-sectional area of the core, which are
given in Table 4.1. Considering duty loss of 20% at the secondary winding of the transformer
originated from the leakage inductance, the final number of turns for primary and secondary
windings are determined to be,
Np : Ns = 6 : 90
d. Wire size
The wire size of the transformer windings is calculated based on the rms value of the winding
current. Since the winding current is 2 times larger than the switch current at each side of the
transformer, the rms vales of the primary and secondary winding currents can be calculated to
-
20
be 187A and 12.5A, respectively. Then, at 500 circular mils per rms ampere the required number
of circular mils is obtained by [9],
Primary circular mil requirement 500,93500 == rmsI
Secondary circular mil requirement 900,5500 == rmsI
Hence, the wire sizes AWG 0 and AWG 12 are selected from AWG table for the primary and
secondary wires, respectively.
4.1.2 Control Method
Fig. 4.4 shows the block diagram for the feedback control of the front end DC-DC
converter. The first goal of the control is to regulate the dc link voltage. A PI compensator
is used for the voltage control. A current control is also implemented to improve the
dynamic characteristic of the system and to reduce current ratings of the power components
during load transient condition.
The current reference is restricted by a current limiter whose value is adjusted by a signal
from the DSP based on the fuel cell current command so that the power drawn from the fuel
cell does not exceed its capability. A low cost phase shift PWM controller UCC3895 is
employed for control of the front-end DC-DC converter. The UCC3895 implements control
of a full-bridge power stage by phase shifting the switching of one half-bridge with respect
to the other. It allows constant frequency PWM in conjunction with resonant zero-voltage
switching to provide high efficiency at high frequencies. A useful feature provided by the
UCC3895 is soft start capability that allows the system to be protected from capacitor
inrush currents.
Voltage Controller
CurrentController
Front endDC-DC Converter
Current limiter
FCCCurrent Command
Fuel CellDC Power Input
To InvCurrent
Ref.
DC link Voltage
Fuel Cell Current
Voltage Ref.
Fig. 4.4 Control block diagram for front end DC-DC converter
-
21
4.1.3 Sensing & Protection
The front-end DC-DC inverter provides the protection capability of over current, over/under
voltage and over temperature in the circuit. The UCC3895 provides the capability to detect any
fault signal through an input pin of the chip and will shut down the chip by disabling all the gate
signals to the front-end DC-DC converter.
This shutdown process is accomplished by any one of the following conditions : 1) the dc
link voltage measured exceeds a threshold voltage of 500V. 2) the fuel cell voltage goes over
41V or under 22V. 3) the temperature of a bimetal that is mounted on the heat sink of the DC-
DC converter rises over 80C. 4) the current drawn from the fuel cell exceeds 100% of the
maximum fuel cell current during longer than 1 minute . A resistive divider followed by an
isolation amplifier has been used as a voltage sensing circuit. The fuel cell current is sensed
through a low resistance shunt resistor for over current protection. Fast fuses have been
used to protect the DC-DC converter itself from being damaged by any fault of the other
section of the inverter system. The schematic diagram for the sensing circuit is shown in the
Appendix A.1.
4.2 DC-AC Inverter
The DC-to-AC inverter section, shown in Fig. 4.5, is located between the DC-DC converter
and the load.
SW3 SW1
SW2SW4
C1
C2Cf Cf
Lf
Lf
400VDC
ISA Ia
Ib VabA
B
N
VaVb
IC1
Fig. 4.5 Circuit diagram of the inverter
-
22
The inverter system consists of two half-bridge inverters, utilizing center tapped dc link
capacitors followed by output filters. The front-end DC-DC converter maintains equal 200V
on the dc-link capacitors, and two inverter legs are operated to generate a split single-phase
120/240Vac, 60Hz output.
A low cost DSP is implemented to provide the control scheme for the inverter system. A
digital PI compensator is employed to regulate the output voltage under varying load condition.
4.2.1 Inverter design
In this section the ratings of the power components in the inverter system are determined. A
detailed list of the inverter requirement for our inverter design is,
- 5kW continuous @ displacement factor 0.7 leading or lagging, 10kW overload for 1 min.
- Output voltage : 120/240V(split-phase).
- Output frequency : 60Hz0.1Hz.
- Output voltage THD : less than 5% when supplying a stand nonlinear test load.
- Output voltage regulation quality : output voltage tolerance no wider than 6%.
DC Link Capacitors
For a worst case, 10KW overload for 1 minute at displacement factor of 0.7 is considered,
then the output VA becomes,
VAVAout 142807.010000
== (13)
The full load current of each phase is given by,
AI rmsa 5.59120214280
, == (14)
For the sake of simplicity, the output current ia is assumed to consist of only fundamental (Ia,1)
and third harmonic (Ia,3). Further, assuming Ia,3 = 0.7Ia,1 since this is a typical case of a single
phase rectifier type nonlinear load [5],
1,2
3,2
1,. 22.1 aaarmsa IIII =+ (15)
Therefore, the fundamental rms value of each phase output current becomes,
-
23
AIa 77.4822.15.59
1, == (16)
The most dominant component of the DC-link capacitor current ic1 is the fundamental
frequency current, the rms value of which equals,
AII ac 3.2421
1,1,1 = (17)
For a permissible voltage ripple 1cV less than 10% or 20V, capacitance can be obtained by,
FV
IC
c
c
322220602
3.24
1
1,11 =
=
= (18)
The peak voltage rating of the dc link capacitor C1 becomes,
VVVV cDCpeakc 21021
21
1,1 =+= (19)
Based on these designed values, an actual device of 250V 3300uF was selected from a
manufacture.
Inverter switches
The av current rating of an inverter switch can be obtained by,
AI avSA 28, = (20)
The peak voltage rating of each IGBT is 420V which is the peak dc link voltage.
Based on these designed values, IGBTs with 600V 50A rating were selected from a manufacture.
Output filter design
Assuming the switching frequency fs to be 20kHz, the frequency ratio is,
3.3331
==ffn s (21)
An equivalent circuit for output filter design is shown in Fig. 4.6. The transfer function Hn for
the equivalent circuit can be obtained by,
)( 2,,,
cLnLcL
nLc
in
nan XXnjZXnX
ZjXVV
H+
== (22)
-
24
Vin
jnXL
ZL,n-jXcn Va,h
Ia
Fig. 4.6 Equivalent circuit for a LC output filter
The gain of the transfer function at fundamental frequency, H1, approximates unity if
cL XX (23)
Va,n : output voltage harmonic
Vin : input voltage harmonic
Xc : impedance of capacitor
XL : impedance of inductor
ZL,n : impedance of load
n : harmonic due to the switching
As the load impedance ZL,n approaches to infinity, that is, at no load condition the gain at
harmonic frequency,Hn, approximates in the following,
1
12
2
=
=
c
LcL
cn
XXnXXn
XH (24)
Therefore, to satisfy THD requirement of less than 5%, only the switching frequency
component is considered as [5],
045.01
12
c
L
XXn
(25)
To limit the ripple voltage across the filter capacitor generated from the third harmonic load
current, an equivalent circuit is considered as shown in Fig. 4.7. The current flowing through the
filter capacitor is,
-
25
jnXL
Ia,h
Ic-jXc
n Va,h
Fig. 4.7 Equivalent circuit for a non-linear load
ha
LC
Lc I
jhXh
jXjhXI ,+
= (26)
Then, the voltage across the filter capacitor at a harmonic frequency becomes,
Va,h : equivalent voltage
Ih : current at harmonic
Xc : impedance of capacitor
XL : impedance of inductor
h : harmonic due to non-linear load
ha
C
L
Lha I
XXh
jhXV ,2
,
1
= (27)
This can further approximate,
haLha IhXV ,, (28)
if 12 C
L
XXh (29)
For the third harmonic h = 3,
1,
3,
1,
3, 3
a
aL
a
a
VIX
VV
(30)
The capacitor ripple voltage at the third harmonic frequency is limited to 3% of the fundamental
output voltage. Then, the impedance of the filter inductor can be determined by,
( ) 035.014.34312003.0
303.0
3,
1, =
=
a
aL I
VX (31)
-
26
Then, the filter inductance becomes,
Hf
XL Lf 84.92
602035.0
2 1=
== (32)
From eqns. (25) and (31) the impedance of the capacitor can be obtained by,
XC = 167.46 (33)
Then, the filter capacitance becomes,
FXf
Cc
f 16
21
1
=
= (34)
Based on these designed values, inductors with 100H and capacitor with 20F were selected
from a manufacture.
4.2.2 Inverter control by using a DSP
The control for the entire SNUT inverter system is done with the Texas Instrument
TMS320LF2407 DSP [7]. The DSP is a 40MIPS, fixed-point processor. This chip has 16 PWM
signals, 41 general purpose digital I/O pins, 16 high-speed A/D converter inputs, and a serial
communication port. By implementing the control via DSP, the proposed inverter system will
offer increased flexibility and will minimize component cost. The goal of the DSP control is as
follows: 1) the PWM gating signals for IGBTs in the inverter stage are generated according to
the modulation index 2) all of the sensing parameters are sent back to the DSP and are
monitored for control and for fault conditions 3) output voltage regulation is implemented to
meet THD specifications under varying load conditions 4) communication between the DSP in
the inverter system and the fuel cell controller are provided through RS485 and two TTL signals
5) The current reference for the bi-directional DC-DC converter is calculated by comparing the
fuel cell current command to the output real power, and the resultant reference with
charge/discharge mode is sent to the bi-directional DC-DC converter.
4.2.3 Voltage regulation method
Output voltage tolerance should not be wider than 6% over the full line voltage and
temperature range, from no-load to full-load. To meet the output voltage tolerance requirement
-
27
the AC output voltage is sensed and a closed-loop control is implemented with a digital PI
compensator in the DSP.
The AC output voltage sensing circuit consists of a potential transformer (PT), a gain stage, an
offset stage and a filtering stage. The PT is low in cost and has an isolation. The output of the
gain stage is sent to the offset stage, which is necessary because the A/D converter in DSP is
unipolar. The last stage is a high frequency noise filter which is the unity gain, non-inverting 2th
order Butterworth with cutoff frequency of 5kHz. The output voltage from the sensing circuit is
fed to the DSP and is subtracted from a sine wave reference. This error signal is applied to a PI
compensator and then the resultant signal is compared to a triangular wave of 20KHz. A
sinusoidal PWM signal is generated and sent to the gate drive circuit for IGBTs. All the PI
compensation and the sinusoidal PWM generation are implemented within the DSP. Thus, the
DSP will adjust the modulation index to keep the output voltage regulated under unbalanced
load condition and from no-load to full-load. The circuit diagram for the DSP board and the gate
driver are shown in the appendix A.3 and A.4.
4.2.4 Capacitor voltages balancing
Unbalance in dc-link capacitor voltages causes generation of even harmonics in the inverter
output voltages. A control method of balancing the capacitor voltages is shown in Fig. 4.8.
Va.dc LPF
*Va.dc = 0
*
*
Vb
Va
Vb
Va PWM.A
PWM.B
Phase.A offset
INVP
-+
+-
-+ +-
Fig. 4.8 Control method of balancing the capacitor voltages.
-
28
Suppose output voltage Va has a positive dc offset, which means that the upper capacitor voltage
is greater than the lower capacitor voltage. The output voltage Va is sensed and passed through a
low pass filter to obtain a dc component of voltage Va. This causes addition of a positive value
to the reference output voltage Vb* resulting in a decrease in upper capacitor voltage and an
increase in lower capacitor voltage.
4.2.5 Sensing and protection
The SNUT DC-to-AC inverter provides the protection capability of over-current, short circuit,
and over temperature in the circuit to prevent damage to the front-end DC-DC converter stage,
fuel cell and inverter itself. Over-current protection is implement by using a current transformer
followed by several op-amp stages. If the output current measured ranges between 100% and
110% of full load current over 1 minute, a signal is sent to the gate drive for shutdown and to
the DSP to light up the over current fault LED. Also, the DC-to-AC inverter can be protected
from output short circuit. If the output current measured exceeds a threshold of 110% of full
load current, a signal is sent to the gate drive for immediate shutdown and to the DSP to light up
the short circuit fault LED. Temperature protection is implemented by using a bimetal as a
temperature sensor that is mounted on the heat sink of the inverter. If the temperature of the
sensor rises over 60C a fan on the heat sink starts to operate. If the temperature of the sensor
rises over 80C a signal is sent to the gate drive for immediate shutdown and to the DSP to light
up the over temperature fault LED. The inverter also shuts down for safe operation if the DC
link voltage goes over 500V or under 300V. The schematic diagram for the sensing circuit is
shown in the Appendix A.2.
4.3 Bi-directional DC-DC converter
4.3.1 Description and Approach
The fuel cell has a slow response, and therefore power demand from the load and power
supply from the fuel cell does not coincide during the transient load condition.
-
29
Fig. 4.9 Bi-directional DC-DC converter
Therefore, a secondary energy source is required to match the power difference between the
fuel cell and the load. The SNUT team decided to connect the 48V lead acid battery pack that
will be provided at the competition site into the 400V dc link of the SNUT inverter system.
Therefore, a bi-directional DC-DC converter is employed to charge or discharge the battery.
High voltage batteries could be directly connected to the 400V dc link without any intermediate
power converter, but the high voltage battery is relatively expensive and may have the battery
cell unbalance problem. The power converter topology of choice between the battery and the dc
link is the high frequency bi-directional DC-DC converter as shown in Fig. 4.9. The current-
source push-pull converter on the battery side is operated to discharge the battery whereas the
voltage-source full-bridge converter on the dc link side is operated to charge the battery. The
push-pull converter employs MOSFETs as switching devices due to its low voltage and high
current operation whereas the full-bridge converter employs IGBTs due to its high voltage and
low current operation.
According to the recently changed battery management policy by FCT, the battery
management is performed by the fuel cell system.
The DSP in the inverter system is required to determine current reference for the bi-
directional DC-DC converter by calculating the difference in real power between the fuel cell
system and the load.
-
30
Fig. 4.10 Control block diagram for the bi-directional DC-DC converter
Then, the bi-directional DC-DC converter is operated to charge or discharge the battery
according to the current reference and the mode of operation determined by the DSP. The two
PWM controllers, UCC3825 and UCC3895, are employed for charge and discharge modes of
operation, respectively. The control block diagram for the bi-directional DC-DC converter is
shown in Fig. 4.10. The PI control is implemented for current control of both converters. The
inductor current is sensed and compared to a reference determined by the DSP, and then the
error is applied to a PI compensator.
The PWM controller generates the gating signal for switching devices of a converter. Only
one of the two converters should be in operation while the other is in idle state. If the current
drawn from the battery exceeds 120A which is the maximum discharging current, the chip will
shut down the bi-directional DC-DC converter by disabling all the gate signals. The battery
voltage is fed to the DSP and is used for the calculation of the charging current reference. When
the battery is in deep discharge (< 42V) or over charge (>56.7V) states, the DSP will shut down
the bi-directional DC-DC converter to protect the battery from being damaged.
Fig. 4.11 shows the inductor voltage and current waveforms for charge and discharge modes,
respectively. Let us define the turns ratio n2 of the high frequency transformer T2 to be,
p
s
NNn =2 (35)
-
31
(a) Charge mode (b) Discharge mode
Fig. 4.11 Inductor voltage & current waveforms
During the charge mode, we have
SbattSdc
batt TDVDTnVV )
21()(
2
= (36)
which gives
2
2nD
VV
dc
batt = (where, 0 < D < 0.5) (37)
During the discharge mode, we have
Sddc
battSdbatt TDnVVTDV )
21()(
2
= (38)
which gives
)21(2
dbatt
dc
Dn
VV
= (0 < Dd < 0.5) (39)
)1(22
Dn
VV
batt
dc
= (0.5 < D < 1, where D = 0.5+Dd) (40)
4.3.2 Power component design
In this section the power component ratings of the bi-directional DC-DC converter is
determined. The system parameters for the design are given in the following.
The switching frequency for both modes are assumed to be 20kHz
-
32
Battery voltage Vbatt : 48V nominal ( 42V~56.7V)
DC link voltage Vdc : 400V nominal ( 380V~420V assuming 10% of ripple)
Output power at the dc link Pdc = 5000W (max. 1 min.)
Turns ratio of the high frequency transformer
It can be seen from Fig. 4.13 that
02
-
33
directional DC-DC converter should be able to supply full power of 5KW at a maximum load of
10KW. The maximum dc current at the output can be calculated as,
AVPI
dc
dcdc 5.12== (46)
Ignoring power loss in the bi-directional DC-DC converter, average power on the battery side is
the same as average power on the dc link side. Then, the average inductor current become,
)1(22
DInI dcL
= (47)
Then, the average inductor current at a maximum discharge becomes,
IL = 113.6 (A) when D = 0.67.
When switches S1 and S2 are turned on during the discharge mode we have (See Fig. 4.14(b)),
Sd
Lbatt
TDi
LV
= (48)
Then, combining eqn. (48) and D=0.5+Dd the ripple current can be determined by,
batts
L VfLDi
=)5.0( (49)
From eqns. (47) and (49), the rms current rating of the inductor becomes
AI rmsL 113= (50)
Therefore, we choose an inductor with the inductance of 39uH and the rms inductor current
rating of 113A.
Switch ratings
Since the maximum battery discharge current is much larger than the maximum battery
charge current, the switch ratings of the bi-directional DC-DC converter should also be
determined based on the discharge mode of operation at full load (5KW, 1min.).
The voltage and current ratings at the worst case are listed in Table 4.2. Actual devices have
been selected from a manufacture.
-
34
Table 4.2 Voltage and current ratings of the switch based on maximum discharge operation
Devices Current rating Voltage rating Actual device selected
MOSFETs
S1,S2
Ipeak = 119.6A
Iav = 60A Vpeak = 140V
200V. 100A
MOSFETs
IGBTs
S3~S6
Ipeak = 19.93A
Iav = 9.5A Vpeak = 420V
600V. 20A
IGBTs
Transformer Design
In order to reduce the leakage inductances of the high frequency transformer two 2.5KW
transformers are connected in parallel to form a 5KW transformer. The reduced transformer
leakage inductance s results in reducing the rating of the snubber mounted on the primary side.
The 2.5KW transformer design is described in the following.
a. Transformer core material
Ferrite core is chosen as a material of a high frequency transformer due to its low cost
and low losses characteristics for transformers and inductors in the frequency range 20 KHz
to 3MHz.
b. Transformer core size
The SNUT team would follow the procedure for transformer core size selection provided by
Magnetic Inc. [8]. The core type of choice is EER core, then
AmpcmC /1007.5 23= (51)
The transformer efficiency is assumed to be 90%. The winding factor is K = 0.3(primary side
only). The flux density B is assumed be 2000(gauss) and fs = 20KHz.
Then, from eqn(9) the WaAc product is given by,
483
34.293.02000020009.04
10)1007.5(2500 cmAW ca =
=
(52)
Using the core selection table by area product distribution, a core of 47054-EC(MAGNETICS
Inc.) was selected. Magnetic data for the selected core is shown in Table 4.1.
c. number of turns
The number of primary turns is given by [8],
-
35
turnsBAf
VNs
Pp 81.252000039.320004
101404
10 88=
=
= (53)
From the turns ratio which has been obtained by n2=6, the final number of turns for primary and
secondary windings are determined to be,
Np : Ns = 26: 156
d. wire size
At a discharge mode of operation at the full load (5KW, 1min.), the rms vales of the primary
and secondary winding currents are 39.2A and 8.4A, respectively. Then, at 500 circular mils per
rms ampere the required number of circular mils is obtained by [9],
Primary circular mil requirement = 600,19500 = rmsI
Secondary circular mil requirement = 200,4500 = rmsI
Hence, the wire sizes AWG 7 and AWG 13 are selected from AWG table for the primary and
secondary wires, respectively.
4.4 System interface
4.4.1 Communication interface
The SNUT team will connect a PC into the inverter system so that the output phase
voltage, the output phase current, the output power, the frequency of the output voltage and
some inverter status including faults, etc. could be monitored. All the data monitored are
also recorded in the PC and updated every two minutes so that inverter status could be
interpreted after the fault has occurred. The programming language, Visual C++, has been
used for PC monitoring program.
4.4.2 Monitoring Function
Fig. 4.13 shows an example view of the PC monitor display. The explanation on the display
menu is given in the following.
Inverter System Status
Input Voltage : Display actual input voltage of the front-end DC-DC converter.
-
36
Fig 4.12 Display of RS-232
Input Current : Display actual input current of the front-end DC-DC converter.
Input Power : Display actual input power of the front end DC-DC converter.
Output Voltage : Display actual output voltage between phases A and B of the inverter.
Output Current : Display actual output current between phases A and B of the inverter
Output Power : Display actual output average power between phases A and B of the
inverter
OL time : Display actual over load time.
DC Link Voltage : Display actual dc-link voltage.
Bi-direction System Status
D_charge ref. : Display the current reference for battery discharge
Charge ref. : Display the current reference for battery charge.
Battery Voltage : Display actual battery voltage.
-
37
Inverter Output : Phase A / B
Frequency : Display actual frequency of the Inverter output voltage.
Voltage : Display actual inverter output voltage in rms.
Current : Display actual inverter output current in rms.
V_mean : Display actual inverter output voltage offset.
Power_avg : Display actual real power(W) at the inverter output.
Power_app : Display actual apparent power(VA) at the inverter output.
P_factor : Display actual power factor at the inverter output.
Fault Status
Over Load (5kw for a minute) : Over Load LED
(when the over load continue for a minute)
Under Voltage Trip (21V) : Under Voltage Trip LED
(when the input voltage is under 21V)
DC Link Fault (higher than 480V) : DC Link Fault LED
(when the DC-DC converter Output is higher than 480V)
DC Link Fault (lower than 300V) : DC Link Fault LED
(when the DC-DC converter Output is lower than 300V)
Low Battery Fault (lower than 48v) : Low Battery Fault LED
(when the battery voltage is under than 48V)
4.5 Heat sink analysis
4.5.1 Thermal analysis
The heat generated by electrical loses from the switching devices should be dissipated to
avoid the performance degradation or failure. The heat sink is a crucial and a costly
component of the power processing unit. The factors to be considered for designing the heat
sink are material, weight, size, maximum heat load, surrounding temperature and cost. A
fan will be employed to increase the rate of convection heat transfer rate of the heat sink
-
38
making the size of the heat sink smaller. In this section the design of the heat sink for the
SNUT inverter system is detailed. The operating parameters such as total power dissipation
are defined and thermal circuits of the switching devices mounted on the heat sink are
established and analyzed. The first step is to calculate the power dissipation of a switching
device according to the equations below.
MOSFETs
Switching loss ][21
)()( offonspeakdcrmsdssw ttfVIP += (54)
Conduction loss s
ononDSrmsSWcon t
tRIP = )()(2 (55)
Total loss swcon PPP += (56)
IGBTs
Switching loss ][21
)()( offonsrmsCEpeakdcsw ttfIVP += (57)
Conduction loss s
onrmsCEsatCEcon t
tIVP = )()( (58)
Total loss swcon PPP += (59)
Then, a thermal equivalent circuit for analyzing thermal characteristic of the heat sink is defined
as shown in Fig. 4.13 when two kinds of power devices are mounted on a heat sink.
Given power loss Pl (where, l = 1 or 2) of a switching device, junction to case thermal
resistance Rjc,l case to heat sink thermal resistance Rch,l ambient temperature Ta, and juntion
temperature Tj,l heat sink to ambient thermal resistance Rha can be obtained in the following
procedure.
The case temperature Tc,l can be given as,
ljclljlc RPTT ,,, = (60)
-
39
Fig. 4.13 Thermal equivalent circuit
Then, heat sink temperature Th,l can be given as,
lchllclh RPTT ,,, = (61)
Then, the total heat sink temperature Th is,
2,1, hhh TTT += (62)
Finally, heat sink to ambient thermal resistance Rha is obtained by,
21 PPTTR ahha +
= (63)
4.5.2 Heat sink design
In this section the heat sink for SNUT inverter system is designed based on the actual devices
selected in the previous section. The switching loss, the conduction loss and the total power loss
of MOSFETs and IGBTs in each section are calculated using eqns. (54) to (59) and listed in
Table 4.3. A heat sink will be used for each section of the converters.
Now, the ambient temperature Ta, which is the temperature inside the package, is assumed to
be 40C. The case to heat sink thermal resistance Rch is usually considered to be 0.3C/W if a
thermal grease is applied between the case and the heat sink. The junction temperature Tj is
given in the data sheet of the power device. Then, the case temperature Tc, the heat sink
temperature Th and the heat sink to ambient thermal resistance Rha can be obtained using eqns.
(60) to (63) Table 4.4 summarizes the thermal characteristics required for the heat sink design.
Using the heat sink to ambient thermal resistance, the area of the heat sink required can be
-
40
calculated or the heat sink can directly be selected from a manufacture by the heat sink to
ambient thermal resistance obtained. Fig.4.14 to Fig.4.15 shows the heat sinks of each section
selected based on the heat sink to ambient thermal resistance Rha. A safety margin was
considered for actual heat sink selection from the manufacture.
Table 4.3 Power dissipation in the device used
Section Device Switching loss per unit
(Psw) Conduction loss per unit
(Pcon) Total loss per unit
(P)
MOSFET 16.7 32.2 48.9 Front-end DC-DC Diode 0.012 13.2 13.21
Inverter IGBT 70.56 50.4 120.96
MOSFET 48.3 59.4 107.7 Bi-directional DC-DC IGBT 34.9 5 39.9
Table 4.4 Thermal characteristics for the heat sink design
Section Device Q`ty Rjc(C/W) Tj(C) Tc(C) Th(C) Rha(C/W)
MOSFET 12 0.21 70.26 60 Front-end DC-DC Diode 4 1.25 65.85 49.34
45.38 0.11
Inverter IGBT 4 0.22 112.5 85.9
MOSFET 2 0.18 103.3 80.92 Bi-directional DC-DC
IGBT 4 0.31 67.54 55.17
49.7 0.08
Fig. 4.14 Heat sink for front-end DC-DC converter (Rha=0.09C/W)
-
41
Fig. 4.15 Heat sink for inverter and bi-directional (Rha=0.08C/W)
5. Simulation
In this section the whole inverter system designed in the previous section is simulated using
PSIM to validate the proposed design concept. The simulation was done with realistic
parameters of the selected device if possible. Fig. 5.1 shows the simulated waveforms for a
worst case of load transient from 5KW to 10KW and back to 5KW again. Fig. 5.1(a) shows the
inverter current on the dc side demonstrating an increase and a decrease in average value
according to the load change. Fig. 5.1(b) shows the output current of the front-end DC-DC
converter. The average value of the current did not change during the transients even if the
ripple was increased due to the operation of bi-directional DC-DC converter. The dc link
voltage was well regulated at 400V, as shown in Fig. 5.1(c). Fig. 5.1(d) and (e) shows the
current waveforms of the dc link side and the battery side of the bi-directional DC-DC converter,
respectively. This illustrates that for a sudden load change the bi-directional DC-DC converter
draws an amount of power from the battery, which is difference in power between the fuel cell
and the load. The output phase voltage was well regulated during the load transient while the
output current shows an increase and a decrease according to the load change.
-
42
Fig. 5.1 Simulated waveforms (5KW 10KW 5KW) (a) inverter current on the dc side, (b) output current of the front-end DC-DC converter, (c) dc link voltage, (d) current waveforms on the dc link side of the bi-directional DC-DC converter, (e) current waveforms of the battery side of the bi-directional DC-DC converter, (f) output phase voltage, (g) output phase current
(a)
(b)
(c)
(d)
(e)
(f)
(g)
-
43
6. Experimental Result
A 10KW prototype inverter has been built in the laboratory of SNUT, and experimental
waveforms are presented in this section. A programmable DC power source capable of
supplying 5KW was used instead of the fuel cell. Four 12V, 80AH batteries are connected in
series to form a 48V battery. The experimental waveforms for a steady state condition at
4.4KW load level are shown in Fig. 6.1. Fig. 6.1(a) shows the output phase voltages VAN and
VBN, respectively. Fig. 6.1(b) shows the output voltage VAB and current with a linear load.
The experimental waveforms were obtained for a transient discharge mode of operation, that
is, a load increase from 2kW to 2.7kW. The upper trace in Fig. 6.2(a) shows the output current
of the front end DC-DC converter whose average value did not change after the transient even if
the ripple was slightly increased due to operation of bi-directional DC-DC converter. The lower
trace in Fig. 6.2(a) shows the dc link voltage, which undergoes an overshoot and is stabilized.
Fig. 6.2(b) shows the inverter current on the dc side indicating an increase in average value
according to the load increase. Fig.6.2(c) and (d) show the PWM current waveform on the dc
link side of the bi-directional DC-DC converter and its extended waveform in time scale,
respectively. Fig. 6.2(e) and (f) show the current waveform on the battery side of the bi-
directional DC-DC converter and its extended waveform in time scale, respectively. This
demonstrates that for a sudden load increase the bi-directional DC-DC converter would quickly
draw an amount of power from the battery, which is difference in power between the fuel cell
and the load. The upper trace in Fig. 6.2(g) shows the output phase voltage which is well
regulated during the load increase. The lower trace in Fig. 6.2(g) shows a magnitude increase of
the output phase current indicating load increase. Fig. 6.2(h) shows the extended waveforms in
time scale for Fig. 6.2(g).
Photograph of the SNUT fuel cell inverter system is shown in Fig. 6.3..
-
44
(a) output voltages : phase AN and phase BN
(b) output voltage and current : phase AB
Fig. 6.1 Experimental waveforms (4.4kW load)
-
45
Fig. 6.2 Experimental waveforms ( 2KW 2.7KW ), (a) upper : output current of the front end converter ; lower: dc link voltage, (b) dc side inverter current, (c) output current of the bi-directional converter, (d) extended waveform of (c), (e) battery current, (f) extended waveform of (e), (g) output phase voltage and current, (h) extended waveforms of (g)
(b) (a)
(c) (d)
(e) (f)
(g) (h)
-
46
Fig. 6.3 Photograph of the SNUT fuel cell inverter
7. Performance evaluation
The 10kW prototype inverter system was tested from no load to 4.4kW load. Experimental
performances of some important design items have been obtained and compared to minimum
target requirement of the inverter system as shown in Table 7.1. The SNUT prototype inverter
met the minimum target requirements for most of the design items such as frequency regulation,
THD of the output voltage, output voltage regulation and input current ripple. The SNUT
prototype demonstrated a good performance, especially in THD of output voltage and output
voltage regulation. The efficiency of the front end DC-DC converter section was 90% and that
of the inverter section was 97% resulting in total system efficiency of 88%.
The SNUT team is trying to increase the efficiency by optimizing design and selection of the
devices.
-
47
Table 7.1 Experimental performance (no load to 4.4kW load)
Design Item 2003 FEC
Specification performance SNUT team
Experimental performance
Frequency 60Hz 0.1Hz 59.95Hz ~ 60.09Hz
THD (Output voltage harmonic) 5% Lower than 1.94%
Regulation 6% -2.4% ~ +0.2% Input current ripple 3% Lower than 2.2%
Efficiency (measured at 4.4kW load) Higher than 90% Total 88%
(DC-DC:90%, INV:97%)
8. Bill of Materials
A detailed bill of materials for the front end DC-DC converter, the DC-AC inverter, and the
bi-directional DC-DC converter sections is listed in Table 8.1. We could not find some electrical
parts such as power switching devices and transformer cores in Korea. Therefore, we sometimes
had to use parts which has much larger ratings than the designed value.
9. Cost analysis
The SNUT team has been placing great emphasis on cost through the whole design process.
There are many factors to purchasing electrical parts. The cost analysis is based on the
spreadsheets evaluation forms provided in the 2003 FEC workshop. The result of the cost
analysis for the front end DC-DC converter, the DC-AC inverter and the bi-directional DC-DC
converter are shown in Table 9.1 to 9.3, respectively. The cost of the front end DC-DC converter
was $233.23. The cost of the DC-AC inverter was $150.06. The cost of the bi-directional DC-
DC converter was $121.1. The total cost of the 10kW SNUT fuel cell inverter system was
$504.39. The values in the table indicate only preliminary, relative cost estimates, not dollars. A
detailed bill of material will be developed and provided in the final report for evaluation of
actual cost of product.
-
48
Table 8.1 Bill of materials
Component (Rating) Manufacturer (Part number) Qty.
Power switch MOSFETs (100V, 180A) IXYS (IXFN180N10) 12 MOSFETs (200V, 180A ) IXYS (IXFN180N20) 2 IGBTs (600V, 200A ) Fuji (2MBI200N-060) 2 IGBTs (600V, 50A) Toshiba (MG50J2YS50) 2 Diodes (1000V, 2X30A ) IXYS (DESI 2X31-10B) 5
Inductor MPP core (100uH, 15A) Chang-Sung (CH270125E) 2 MPP core (93uH, 60A) Chang-Sung (CH572060E) 2 MPP core (40uH, 120A) Magnetics (K5528B026) 1
Transformer Ferrite EE core (2.5kW) ISU ceramics (EE118) 4
Capacitor Electrolytic (400V, 1000uF) 2 Electrolytic (400V, 3300uF) 2
Opto-isolated gate driver IC Agilent (HCPL-316J) 14 Control
DSP controller TI (TMS320LF2407) 1 Unitrode (UCC3895) 2
PWM controller Unitrode (UC3825) 1
Amplifier Isolation amplifier Burr-Brown (ISO122) 2 Instrumentation amplifier Burr-Brown (INA126) 2 Differential amplifier Burr-Brown (INA117) 1 Op-amp National (LF353) 20
Fans (120Vac, 33W) Fulltech (UF15P-12H) 3 Sensor
Hall current (400A) LEM (HAS 400S) 1 Hall current (200A) LEM (HAS 200S) 3
Table 9.1 Cost spread sheet for front end DC-DC converter
VOLT VOLT CUR CUR UNIT EXTENDEDDEVICE QTY DESIG UNIT MEASURE (Vpk) (Vrms) (Avg) (Arms) COST COST
DIODE 8 D1~D8 410 6.25 2.29 18.34
MOSFET 4 S1~S4 41 125 6.56 26.24
CAP (ALUM) 2 C1,2 3222 uF 210 19.78 39.55
TRANSFORMER 1 T1 47.3 389.8 26.57 26.57
CHOKE 2 L1, 2 100 UH 12.6 41.40 82.79
LOSSES 416.8 W 34.73 34.73
CONTROL 5.00
TOTAL 233.23
-
49
Table 9.2 Cost spread sheet for inverter
Table 9.3 Cost spread sheet for bi-directional converter
10. Conclusion
The objective of this project is to develop a low cost, high efficiency 10kW inverter system
for a SOFC system. In this report a power circuit topology for the inverter system has been
chosen after evaluating two possible topologies in a practical way proposed. All the component
ratings were designed along with thorough analysis on the chosen topology. A hardware
prototype capable of supplying 10kW load was built and tested at a laboratory of Seoul National
University of Technology. The SNUT prototype inverter met the minimum target requirements
and demonstrated a good performance in most of the design items. The SNUT team has been
trying to increase the efficiency and to decrease the cost by optimizing design and selection of
the devices. The SNUT team strongly believes the final prototype meet the efficiency and cost
requirements.
VOLT VOLT CUR CUR UNIT EXTENDEDDEVICE QTY DESIG UNIT MEASURE (Vpk) (Vrms) (Avg) (Arms) COST COST
IGBT 2 S1~S4 420 28 4.77 9.53
CAP (ALUM) 2 C3,4 16 uF 170 0.17 0.33
CHOKE 2 L3,4 93 UH 60 59.50 119.00
LOSSES 134.4 W 11.20 11.20
CONTROL 10.00
TOTAL 150.06
VOLT VOLT CUR CUR UNIT EXTENDEDDEVICE QTY DESIG UNIT MEASURE (Vpk) (Vrms) (Avg) (Arms) COST COST
IGBT 2 S11~S14 420 9.5 1.69 3.38
MOSFET 2 S9, S10 140 60 8.32 16.65
CHOKE 1 L5 40 UH 113 69.49 69.49
TRANSFORMER 1 T2 58 73.3 10.95 10.95
LOSSES 151.5 W 12.63 12.63
CONTROL 8.00
TOTAL 121.10
-
50
11. Reference
[1] R. Anahara, S. Yokokawa and M. Sakurai, Present Status and Future Prospects for Fuel Cell Power
Systems, Proceedings of the IEEE, vol. 81, no. 3, March 1993, pp.399-407
[2] A. Emadi and S. Williamson, Status Review of Power Electronic Converters for Fuel Cell
Applications, Journal of Power Electronics, vol. 1, no. 2, Oct. 2001, pp.133-144
[3] N. Azli, A. Yatim, DSP-based Online Optimal PWM Multilevel Control for Fuel Cell Power
Conditioning Systems, IEEE IECON conf. rec, 2001, pp.921-926
[4] Final Reports from the 2001 Future Energy Challenge, Available: http://www.energychallenge.org
[5] R. Gopinath, D. Kim, J. H. Hahn, M. Webster, J. Burghardt, S. Campbell, D. Becker, P. N. Enjeti, M.
Yeary, J. Howze, Development of a Low Cost Fuel Cell Inverter System with DSP Control, Power
Electronics Specialists Conference, 2002. pesc 02. 2002 IEEE 33rd Annual , Volume: 1 , 2002, pp. 309-
314
[6] The 2003 International Future Energy Challenge
http://www.energychallenge.org/
[7] Texas Instruments, http://www.ti.com/
[8] Design application note MAGNETICS. INC.
[9] A. I. Pressman, Switching Power Supply Design. McGRAW-HILL INTERNATIONAL, 1999.
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51
Appendices
Appendix A.1 Schematic for the sensing board
1M
0.1u
10n0.1u
0.1u
0.1u
DC-DC converter(15v)
1
2
5
3
4Vin
GND
Vout(+)
GND
Vout(-)0.1u
200k
10k
ISO122
1
2
7
89
10
15
16
+Vs1
-Vs1
Vout
GND+Vs2
-Vs2
Vin
GND
V_Sensing
0.1u
0.1u
INA 126
7
3
8
2
6
54
1
V+
Vin+
-
Vin-
Vo
GNDV-
-
10k
SMPS(+15V)
SMPS(+15V)
0.1u
0.1u
10n
SMPS(-15V)
Vout DC 400V
DC-DC(+15V)
DC-DC(-15V)1k 10k
200k
DC-DC(+15V)
DC-DC(-15V)
Appendix A.2 Schematic for the sensing and protection
+15
+5
-
+
U8ALF353
3
21
84
+15
-15
-15
OV_Ian
T1
TRAN_HM31
1 3
2 5
ON : SD
+15
+15
0.1uF
360
100
D5
1N4148
Vbn_6V
R3
20k
Van
-15
101
10k
-
+
U3BLF353
5
67
84
10k
0.1uF
Ibn_rms
-
+
U4BLF353
5
67
84
R3
20k
201
+15
Ian
OV_Ibn
10k
10k
G1SG1
-
+
U3ALF353
3
21
84
U7B
4081
5
64
147
10k
+15
-
+
U1BLF353
5
67
84
0.1uF
-
+
U8BLF353
5
67
84
10k
-
+
U2ALF353
3
21
84
T1
TRAN_HM31
1 3
2 5
10k
OV_I SET
0.1uF
-
+
U1BLF353
5
67
84
101
+15
U11
EL25P1
1
2
3
+
-
M
50k VR1
+5
100
U7A
4081
1
23
147
SG2
R1
10k
-15
Van_rms
Vbn_rms
PWM1
0.1uF
R2
300
+5
10K
-
+
U4ALF353
3
21
84
10k
-150.1uF0.1uF
+15
10K
Van_6V
10K
D2
DIODE
R2
300
U6D
4081
12
1311
147
U6B
4081
5
64
147
OV_I SET
0.1uF
10k
Ibn_rms
D2
4148
10k
D3
ON :
U7C
4081
8
910
147
0.1uF
+C1847uF
R2
300
0.1uF
10k
G3
0.1uF
-
+
U1ALF353
3
21
84
-15
Vbn
D2
4148
+5
N
R3
1k
20k
Ian_rms
0.1uF
20k
10K
SG4
0.1uF
+5
SG3
D2
DIODE
R3
1k D5
1N4148
SD
-
+
U1ALF353
3
21
84
+C1847uF
D5
1N4148
SW1SD_ON
12
470
U6C
4081
8
910
147
PWM4201
360
Ibn
0.1uF
-15
10k
-15
N
-
+
U2ALF353
3
21
84
U11
EL25P1
1
2
3
+
-
M
10k
470
10k
U8
74HC573
23456789
111
1918171615141312
D0D1D2D3D4D5D6D7
LEOE
Q0Q1Q2Q3Q4Q5Q6Q7
D3
ON : OV_I
D5
1N4148
10k
OFF :
0.1uF
+5
G2
50k VR1
PWM3
10k
+C184.7uF
U6A
4081
1
23
147
+15
10k
G4
D3
ON : OV_I
Ian_rms
0.1uF
+15
10k
PWM2
+C184.7uF
-
52
Appendix A.3 DSP board
-
53
Appendix A.4 Inverter gate driver
0
+5
D11N4746
3.3K0.1uF
0.1uF
100pF
HCPL-316J
1
2
3
4
5
6
7
8 9
10
11
12
13
14
15
16Vin+
Vin-
Vcc1
GND1
RESET
FAULT
V LED1+
V LED1- V EE2
V EE1
Vout
Vc
Vcc2
DESAT
V LED2+
VE
D2
DIODE
G410 1W
DC LINK
0.1uF
Q8
BDW94C
0.1uF
+5
0.1uF
SG4
10 1WFAULT 2
0.1uF
D2
DIODE
0
100pF
D2
DIODE
100pF
10
+15
0.1uF
0.1uF
330pF
Q6
BDW93C
100
10 1W
D11N4746
10 1W
HCPL-316J
1
2
3
4
5
6
7
8 9
10
11
12
13
14
15
16Vin+
Vin-
Vcc1
GND1
RESET
FAULT
V LED1+
V LED1- V EE2
V EE1
Vout
Vc
Vcc2
DESAT
V LED2+
VE
GND_1
Q8
BDW94C
D11N4746
10
Q8
BDW94C
3.3K
100
100pF
3.3K
SG1
10
0.1uF
Q6
BDW93C
100
330pF
+5
0.1uF
GND_1
FAULT 4
D11N4746
0
HCPL-316J
1
2
3
4
5
6
7
8 9
10
11
12
13
14
15
16Vin+
Vin-
Vcc1
GND1
RESET
FAULT
V LED1+
V LED1- V EE2
V EE1
Vout
Vc
Vcc2
DESAT
V LED2+
VE
D11N4746
SG2
FAULT 1
0.1uF
SG3
G2
D2
DIODE
FAULT 3
0GND_1
+5
S3
+15
DC LINK
S1
S2
D11N4746
Q6
BDW93C
47K
Q8
BDW94C
D11N4746
+15
0.1uF
G3G1
0.1uF
47K
GND_1
Q6
BDW93C
HCPL-316J
1
2
3
4
5
6
7
8 9
10
11
12
13
14
15
16Vin+
Vin-
Vcc1
GND1
RESET
FAULT
V LED1+
V LED1- V EE2
V EE1
Vout
Vc
Vcc2
DESAT
V LED2+
VE
-15
47K
+15
-15
-15
10
D11N4746
3.3K
0.1uF
-15
100
0.1uF
DC LINK
0.1uF
330pF
330pF
DC LINK
S4
47K
-
54
Appendix B. Project time line
-
55
Appendix C. Transformer core selection by area product distribution
[Design application notes, MAGNETICS Inc.]
